1. Field of the Invention
This invention relates to a tracking type band-pass filter circuit, and more particularly to improvements in the tracking type band-pass filter circuit operating in a burst mode.
2. Description of the Prior Art
Heretofore, this kind of circuit has widely been employed in a carrier recovery circuit of a demodulator circuit of a digital satellite communication system or the like.
For example, in a 4-phase PSK (Phase Shift Keying) system, modulation is removed by a suitable method from a received modulated wave to obtain a sine wave. This is achieved, for instance, by a method of multiplying the modulated wave four times, a method of using remodulation technique or the like. The sine wave thus obtained contains noise components and, to remove them, the sine wave must be applied to a narrow band-pass filter.
However, the input frequency to the carrier recovery circuit usually fluctuates and this fluctuation sometimes exceeds the band width of a band-pass filter necessary for removing the noise. In such a case, it is desirable to achieve some control to reduce the difference between the input frequency and the center frequency of the band-pass filter to zero so that the input frequency and the center frequency of the pass band agree with each other.
In general, a narrow band-pass filter has the characteristics as shown in FIGS. 1A and 1B. FIG. 1A shows the amplitude characteristic and FIG. 1B the phase characteristic of a typical narrow band-pass filter. The narrow band-pass filter has, in the vicinity of the center frequency f.sub.0 of the pass band, a linear characteristic as shown in FIG. 1B in which it presents a substantially constant phase variation .DELTA..theta. in response to a frequency fluctuation .DELTA.f. By making use of this property, it is possible to detect from the phase difference between the input to the narrow band-pass filter and the output therefrom whether or not the input frequency of the filter is in agreement with the center frequency of the pass band. Further, by the arrangement of a feedback control system for reducing the phase difference to zero at all times, the center frequency of the narrow band-pass filter and the input frequency can be controlled to always agree with each other.
The tracking type band-pass filter circuit is based on the above principle and may comprise either an AFC or an APC system which employ different control systems.
FIG. 2 illustrates in block form the principal part of an embodiment of the tracking type band-pass filter circuit of the AFC system. FIG. 2 indicates a sine-wave input 11, a first mixer 101, a band-pas filter 102, a phase detector 103, a loop filter 104, a voltage controlled oscillator (hereinafter referred to as the VCO) 105, and a second mixer 106. Briefly stated, the operation of the circuit depicted in FIG. 2 is as follows. The sine-wave input 11 is frequency converted by the first mixer 101 and then applied to the band-pass filter 102 to remove noise components. If the frequency f of the input sine wave at a connection point 13 differs from the center frequency f.sub.0 of the band-pass filter 102, a phase difference occurs between the connection points 13 and 14 corresponding to the frequency difference. But by an automatic frequency control (AFC) circuit made up of the phase detector 103, the loop filter 104 and the VCO 105, the VCO oscillation frequency is controlled so that the output voltage from the phase detector 103 may approach zero, and as a consequence, the phase difference resulting from the frequency difference is eliminated; therefore a carrier with no phase error is recovered.
FIG. 3 shows in block form the principal part of an embodiment of the tracking type band-pass filter circuit of the APC system. FIG. 3 indicates a sine-wave input 21, a band-pass filter 201, a loop filter 202, a phase detector 203, a carrier output 22. Briefly stated, the operation of the circuit of FIG. 3 is as follows. The input sine wave 21 is applied to the band-pass filter 201 whose center frequency is variable and in which noise components are removed from the input to provide the carrier output 22. If the frequency of the input sine wave 21 differs from the center frequency f.sub.0 of the band-pass filter 201, a phase difference occurs between the input 21 and the output 22 in accordance with the frequency difference therebetween. But by an automatic phase control (APC) circuit made up of the phase detector 203, the loop filter 202 and the band-pass filter 201 the center frequency of the band-pass filter 201 is so controlled as to make the output from the phase detector 203 approach zero, thereby removing the phase difference based on the frequency difference to recover a carrier with no phase error.
When the tracking type band-pass filter circuit is actuated in a burst mode, the input to the band-pass filter, for example, the signal waveform at the connection point 13 in FIG. 2, becomes a discontinuous sine wave as shown in FIG. 4A. Accordingly, in consideration of the transient response characteristic from the OFF state of the sine wave to the ON state, the tracking type band-pass filter must be designed so that it reaches its steady state rapidly. The time until the phase error of the recovered carrier becomes smaller than a certain allowable value after the start of a burst is defined as the pull-in time t.sub.q. The time t.sub.q is dependent upon the bandwith B of the band-pass filter and, to reduce the time t.sub.q, the band width B must be increased. This is well-known in the art. For increasing the attenuation of the noise components of the input signal, the band width B must be decreased. The high-speed pulling-in and the noise removal are contradictory to each other.
The relationship between the charging and discharging time constants of the loop filter and the pull-in time in the circuit of FIG. 3 is discussed next. Letting .phi. represent the phase difference between the signals at the connection points 13 and 14 or the phase error of the recovered carrier, the phase difference or error is in proportion to a difference f-f.sub.0 between the frequency f of the input sine wave at the connection point 13 and the center frequency f.sub.0 of the band-pass filter 102 in FIG. 2. A control voltage of the VCO 105 in FIG. 2 (whose voltage will hereinafter be referred to simply as the control voltage) is made substantially proportional to the phase difference or error .phi. within an operating range of the circuit. If the control voltage immediately before the burst starts is taken as v.sub.1, and if the feedback loop system is assumed to have reached its steady state at the moment of termination of the burst and if the control voltage at that moment is taken as v.sub.2, the difference, v-v.sub.2 between the control voltage v at an arbitrary moment in the burst and the control voltage v.sub.2 is proportional to .DELTA.V=v.sub.2 -v.sub.1. From the above, it is seen that the difference, .phi.-.phi..sub.2 between the phase .phi..sub.2 in the steady state and the phase .phi. at an arbitrary moment in the burst is in proportion to .DELTA.V=v.sub.2 -v.sub.1. On the other hand it is evident, from the above definition of the pull-in time t.sub.q, that the smaller is the difference .phi.-.phi..sub.2 at a certain moment and consequently the smaller the difference .DELTA.V=v.sub.2 -v.sub.1, the shorter the time t.sub.q becomes.
As such a loop filter, there has heretofore been employed a simple RC circuit as shown in FIG. 5A. If the time constant of this circuit is determined from the condition that its output should reache its steady state by the end of the burst, the variation in the output voltage after termination of the burst also has the same time constant. Accordingly, when the interval between burst is long, even if the steady-state value is reached by a burst signal input, it is discharged before the next burst input is applied, so that charging must start with the initial state in response to the next burst input, and consequently the control voltage difference .DELTA.V cannot be reduced. An increase in the discharge time constant causes an increase in the charge time constant, too, making it impossible to reach the steady state by the end of the burst. As a consequence, in the conventional tracking type band-pass filter circuit, it is impossible to sufficiently shorten the pull-in time t.sub.q and achieve the high-speed pulling-in by a narrow band-pass filter.
In the above description, the control voltage is desired to be made small on the assumption that bursts of the same frequency occur. This case can be dealt with by the use of the loop filter of this invention as shown in FIG. 5B or 5C. However, it is necessary to reduce the control voltage fluctuation not only in such a case but also in the case of a frequency deviation between bursts. In the latter case, it is desirable to minimize the absolute value of the voltage V at the connection point 15 in FIG. 2 with respect to all of the bursts. Since the voltage V is in proportion to the difference between the frequency f of the burst at the connection point 13 and the center frequency f.sub.0 of the filter 102 in FIG. 2, as described to previously, it is possible to put V=A(f-f.sub.0) where A is a proportional constant. Let the maximum and minimum frequencies of the bursts be represented by f.sub.max and f.sub.min, respectively. In the case of using the loop filter shown in FIG. 5B, the input voltage is effective only in the positive polarity, so that to obtain V.gtoreq.0 with respect to all the bursts, it is necessary that EQU A(f.sub.max -f.sub.0).gtoreq.0 (1) EQU A(f.sub.min -f.sub.0).gtoreq.0 (2)
Therefore, a maximum value .vertline.V.vertline.max of .vertline.V.vertline. is given as follows:
.vertline.V.vertline.max=.vertline.A.vertline.(f.sub.max -f.sub.min) (3)
Also in the case of employing the loop filter shown in FIG. 5C, the maximum value .vertline.V.vertline.max is likewise given by the equation (3) for actuating the filter at V.ltoreq.0. If there are no such limitations as represented by either of the equations (1) and (2), it is possible to obtain EQU .vertline.V.vertline.max=.vertline.A.vertline.(f.sub.max -f.sub.min)/2 (4)
by selecting parameters so that f.sub.max +f.sub.min =2f.sub.0. The value given by the equation (4) is half the value of that of equation (3); this is advantageous from the viewpoint of high-speed pulling-in.
Such a method is effective in the TDMA (time division multiplex access) system in which burst signals of different frequencies are input from many stations in one frame period. With the said conventional loop filter, abovesaid effect is not obtained; therefore, the high-speed pulling-in is difficult.
The two methods referred to above are characterized by the use of a loop filter by which the burst voltage is retained (or prevented from rapid attenuation) even after a burst is over, so as to reduce the fluctuation of a control voltage of the AFC circuit. With such an arrangement, however, since tracking of the burst is inevitably affected by an immediately preceding one, the phase difference between the bursts is large when their frequency deviation is large. This invention is intended to eliminate frequency fluctuations of individual bursts by achieving high-speed and high-precision pulling-in even in the case of a large frequency deviation between successive bursts.
This is required, for example, in a satellite communication system in which one satellite station is shared by many earth stations. In the satellite communication system, the frequency fluctuation of the burst signal fall in two classes: one is called an individual frequency fluctuation, which is a frequency fluctuation of each earth station relative to the others owing to differences in the transmitting frequency among the earth stations and the other is a frequency fluctuation resulting from a frequency conversion by a relay station (a satellite transponder in the satellite communication). The latter is common to all bursts of the respective earth stations and hence is called a common frequency fluctuation. In many cases, the common frequency fluctuation is larger than the individual frequency fluctuation. The individual frequency fluctuation occurs instantaneously for each burst, whereas the common frequency fluctuation results mainly from a secular variation of the relay station and hence is far slow compared with the individual frequency variation.
In the past, the tracking type band-pass filter circuits of this kind have been mostly designed without consideration of the distinction between such two frequency fluctuations. In such a case, operating regions such as a pull-in frequency range and so on are limited so as to enable quick pulling-in in response to all frequency fluctuations. With such conventional tracking type band-pass filter circuits, it is impossible to operate satisfactorily for both of a very gentle fluctuation (the common frequency fluctuation) and a quick fluctuation (the individual frequency fluctuation).
To overcome such a defect of the prior art, a method of controlling the control voltage of the AFC circuit by changing over the voltage with a switch in accordance with the presence or absence of a burst has been proposed in U.S. Pat. No. 3,969,678 filed July 3, 1975, entitled "Band Pass Filter Circuit with Automatic Bandwidth Adjust", assigned to the same assignee of the present application. However, this method has the disadvantages of having a complicated circuit structure because of the use of a burst detector, of having the possibility of a malfunction in a poor C/N state, and of a time lag in the operation of the switch.